Method and apparatus for a real time signal integrity oscilloscope

ABSTRACT

A system, method, and apparatus for obtaining a record of logic level transitions within a signal, and for accurately determining a voltage-time pair exhibited by the signal. To achieve these ends, a front-end device may be mated to a real-time sampling system, such as an oscilloscope. The front-end device effectively permits the oscilloscope to observe signals exhibiting greater data rates than otherwise possible without the front-end device.

RELATED APPLICATIONS

This is a Divisional of U.S. application Ser. No. 11/329,962, filed Jan.10, 2006 which claims priority to U.S. Provisional Patent ApplicationNo. 60/642,820, entitled “METHOD AND APPARATUS FOR RECORDING A REAL TIMESIGNAL,” filed Jan. 10, 2005, which are hereby incorporated by referencefor all it teaches.

FIELD OF THE INVENTION

The present invention relates to a method and apparatus for measuring atime varying parameter; more specifically the invention relates to anapparatus for a real time signal threshold event recording and analyzingmethod and apparatus; and more particularly still, the invention relatesto recording a real time signal in order to analyze the jitter of thesignal.

BACKGROUND OF INVENTION

One of the most common tasks involving circuit design relates to thedesign and evaluation of data communication paths. Binary encoded serialdata communication technology has become the predominant form ofinterconnect for many communication standards and is being chosen almostexclusively for new applications that require high data rates. A typicalserial data communication channel employs a data transmission schemethat utilizes a single electrical or optical path over which the data issent. An appropriate data encoding scheme provides for the extraction ofthe data signal at the receiver. The use of a single signal transmissionpath avoids the problems of inter-channel synchronization encounteredwhen using multiple, synchronous data channels.

Although the use of a single signal transmission path avoids themultiple channel synchronization issue, it also forces the data rate ofthat single channel to a maximum. Since the time between symbols isinversely proportional to the data rate, maximum. Since the time betweensymbols is inversely proportional to the data rate, ever increasing datarates cause the time separating data symbols to decrease. Asinter-symbol spacing decreases, the symbols begin to interfere with oneanother causing the determination of the data symbol to becomedifficult, ultimately resulting in bit errors. The presence of noise inthe system compounds the problem of determining the correct value of adata symbol.

The vast majority of current high speed serial data communicationsstandards employ Non Return to Zero (NRZ) encoding with a small minorityutilizing Return to Zero (RZ) and other schemes. NRZ and RZ encoding arebinary encoding schemes. The NRZ encoding scheme is very straightforwardin that a high amplitude represents a “1” value and a low amplituderepresents a “0” value. For RZ encoding a “1” value is represented by apositive going pulse and a low value is represented by a low amplitude.In either of these encoding schemes, the locations of transitions fromone amplitude level to another can give a good estimation of the biterror rate of the system. Since low system bit error rate is the goal ofall data communication systems, it is also the primary metric providedby all serial data communication measurement systems.

The primary types of equipment used to measure the fidelity of serialcommunication systems are oscilloscopes, time interval analyzers and biterror rate test sets. Each of these measurement methods generally relatethe location of the transitions in time relative to the ideal samplingtime along some reference amplitude threshold.

In the case of oscilloscopes, the waveform amplitude is sampled at knowntimes. There are two types of oscilloscopes that are employed for thistype of analysis.

The first type is a real time or digitizing sampling oscilloscope. Thisoscilloscope sequentially samples the waveform at a very high rate(e.g., a rate high enough so that each data symbol is sampled a fewtimes). However, having finite memory, the real time oscilloscope canonly digitize the waveform for a number of samples equal to the memorydepth of the oscilloscope. This results in a real time amplitude recordof the waveform for a period of time. A real time record enables anumber of analysis techniques that take into account the displacement oftransitions relative to an ideal transition location. The idealtransition location is derived from the entire sequence of transitionlocations and can incorporate numerous mathematical techniques toemulate receiver functionality without the need for hardware to emulatethe receiver.

Deficiencies related to real time oscilloscopes have to do with the rateat which they can digitize waveforms. As was stated earlier, at thehighest data rates only a few samples are obtained for each data symbol.This necessitates the need to estimate the transition points based oninterpolation. As will be appreciated, the resulting error is oftentimessignificant. Another shortcoming of real time oscilloscopes is that theyhave bandwidth limitations that are significant relative to the datarates used in serial data communication signals which again result inerrors significant to measurement accuracy.

The second type of oscilloscope is an equivalent time oscilloscope whichmeasures the waveform in relation to a repeating trigger event. Thismeasurement is under-sampled in the sense that only one out of aplurality of triggers is selected for instigation of an amplitudemeasurement. The benefits of this type of measurement are that thesampling hardware can measure the signal with very high bandwidth (e.g.,greater than 60 gigahertz) resulting in very accurate reproduction ofall of the waveform—unlike the real time oscilloscopes which are quitelimited in bandwidth.

One major deficiency of equivalent time measurements is related to theunder-sampling. In an under-sampled measurement technique, themeasurements can only be related to one another via the fact that theywere initiated by similar trigger events. No other information isavailable regarding the relationship of one sample to another.Consequently if an ideal sampling time is desired for analysis purposes,either a reference “clock” signal must be supplied to the oscilloscope,usually in the form of a trigger, or one must be extracted from theincoming waveform with clock recovery hardware. However, even with thepresence of the reference clock signal, the temporal measurement tomeasurement information available with a real time oscilloscope cannotbe produced by the equivalent time oscilloscope. This significantlylimits the types of analyses that are supported by the data, andconsequently limits the types of diagnostic conclusions that can bereached.

Time interval analyzers have circuitry that directly measures thetransition points avoiding the estimation error encountered by real timeoscilloscopes due to interpolation. They are, however, an under-sampledmeasurement system and suffer from similar limitations as the equivalenttime oscilloscopes.

Bit error rate test sets have clock recovery built in and actuallysample the incoming waveform at an estimated ideal sampling time andthreshold. Since the result of this sampling is merely digital and notparametric, only a bit error rate is determined. In order to estimatebit error rate, the sampling point must be moved and consequent biterror rates must be measured at each new sampling point. This method isalso typically very slow relative to other methods.

Shortcomings of the current measurement methods are summarized:

1. Real time oscilloscopes

-   -   a. Systems do not measure threshold transition locations        directly, must estimate them using interpolation.    -   b. Bandwidth limitations relative to data rate reduce accuracy        and limit data rate measurement capability.

2. Equivalent time oscilloscope

-   -   a. Under-sampling results in a non-real time record which cannot        support as rich an analysis as a real time record.    -   b. Low measurement rate, long acquisition time    -   c. Trigger is required, but not always available.    -   d. May require clock recovery.

3. Time interval analyzer

-   -   a. Under-sampling results in a non-real time record which cannot        support as rich an analysis as a real time record.    -   b. Limited amplitude analysis capability    -   c. May require clock recovery.

4. Bit error rate test set

-   -   a. Very limited parametric measurement capability    -   b. Very slow acquisition

In view of the foregoing, there arises a need for an improved thresholdcrossing time measurement and recording method and apparatus. Thefollowing invention addresses and helps to solve and/or minimize theshortcomings of the prior art.

SUMMARY

The invention relates to a real time threshold crossing time measurementsystem which provides a real time record for all transitions occurringover a period of time. While amplitude to time will be described as thetransition event in connection with the various preferred embodimentsdescribed below, the present invention is not limited to thismeasurement or particular measurement. Instead the present invention maybe employed in many other environments in which time varying parametersand functions are recorded for subsequent analysis.

Devices constructed in accordance with the principles of the presentinvention are capable of measuring edge locations at a very high rate(billions of measurements/second) for a period of time limited by thememory depth of the acquisition system. The acquisition and storage ofthe edge locations addresses the real time oscilloscope accuracyproblems associated with the limited bandwidth and interpolation. Italso addresses equivalent time oscilloscope and time interval analyzerlimitations related to under-sampling by providing a real time record.This also avoids the necessity for clock recovery. The speed at whichmeasurements are taken addresses the measurement speed limitationsrelated to the equivalent time oscilloscope and the bit error test set.

Several embodiments are presented that can perform the measurement ofthe present invention. In each of the embodiments, threshold crossingsare converted into known waveforms that are easily measured by a veryhigh speed analog to digital conversion system. The known waveforms arechosen so that the conversion system is able to sufficiently sample soas to infer the time location of the threshold crossing.

An oscilloscope system has also been devised based to implement the timemeasurement system of the present invention. This oscilloscope systemutilizes commodity, high bandwidth, track and hold technology to samplessignals at a rate much higher than equivalent time oscilloscopes. Thetrigger for the sampling system is internal, thereby eliminating theneed for an external trigger. The time measurement system measures thetrigger times at the same time that the amplitude of the waveform isbeing measured—resulting in an under-sampled but real time record. Thiscan all be done at the same time the threshold transition locations arebeing measured as described above. A trigger method that creates anuncorrelated random trigger pattern was conceived that enables extremelyfast measurements that are useful for eye diagram analysis. Two methodsfor enhancing amplitude and timing accuracy have been conceived to allowthis oscilloscope system to meet and or exceed the performancecharacteristics of the equivalent time oscilloscope.

According to some embodiments, a method of determining a time ofoccurrence of a logic level transition in a signal includes providingthe signal to a comparator, so that, at an output of the comparator, astep signal having a substantially predetermined shape is generated,upon the signal exhibiting a logic level transition. The step signal isfiltered, yielding a filtered step signal exhibiting a slowed transitionfrom a first voltage level to a second voltage level. The filtered stepsignal is sampled with a signal measurement device, so as to obtain atleast one sample of the slowed voltage transition. The at least onesample of the slowed voltage transition and the substantiallypredetermined shape of the step signal are used to determine the time ofoccurrence of the logic level transition.

According to some embodiments, a method of sampling a signal includesproviding the signal to a track and hold element. The track and holdelement is caused to perform a hold operation, thereby yielding a heldsignal exhibiting a substantially constant voltage level. The act ofcausing the hold operation includes providing a track signal having asubstantially predetermined shape to the track and hold element. Theheld signal is provided to a signal measurement device. The track signalis provided to the signal measurement device, so that the signalmeasurement device samples the track signal, and obtains at least onesample on an edge of the track signal. The at least one sample and thepredetermined shape of the track signal are used to determine a timevalue associated with the held signal.

According to some embodiments, a method of determining a correctionfactor for a signal obtained by a probe and measured with a measuringsystem includes providing a reference signal to a conductive pad.Coupling the pad with the probe. The measurement system is used tomeasure, in frequency space or time space, the reference signal. Themeasured frequency-space step signal is divided by a frequency-spacerepresentation of the signal provided to the pad, to arrive at thecorrection factor.

It will be appreciated that while the invention is described inconnection with the preferred embodiments in the context of serial datacommunications, the invention can be applied to more general signalanalysis as well. Further, it will be appreciated that the presentinvention need not include each and every one of the preceding features.Instead, methods and apparatus constructed in accordance with theprinciples of the present invention may utilize one or more of theidentified features.

While the invention will be described with respect to preferredembodiment configurations and with respect to particular devices usedtherein, it will be understood that the invention is not to be construedas limited in any manner by either such configuration or componentsdescribed herein. These and other variations of the invention willbecome apparent to those skilled in the art upon a more detaileddescription of the invention.

For a better understanding of the invention, reference should be had tothe drawings which form a part hereof and to the accompanyingdescriptive matter, in which there is illustrated and described apreferred embodiment of the invention.

BRIEF DESCRIPTION OF DRAWINGS

Referring to the drawings, wherein like numerals represent like partsthroughout the several views:

FIG. 1 illustrates a block diagram for one implementation of thereal-time time measurement method of the present invention.

FIG. 2 illustrates waveforms at various points in the time measurementmethod.

FIG. 3 illustrates the block diagram for an alternate time measurementmethod.

FIG. 4 illustrates a block diagram for a dividing scheme that extendsthe scalability of all of the time measurement schemes.

FIG. 5 illustrates the block diagram for an oscilloscope that utilizesthe time previously described time measurement capability to improvesystem timing accuracy and measurement rate.

FIG. 6A illustrates the block diagram for an oscilloscope thatincorporates a randomized trigger source for performing extremely fasteye diagram like oscilloscope measurements.

FIG. 6B depicts an alternative embodiment of a randomized triggersource.

FIG. 7 illustrates a timing loop that is used to reduce the timingmeasurement error noise floor.

FIG. 8 illustrates a likely system configuration for the real timesignal integrity oscilloscope that is capable of measuring amplitude andthreshold crossing timing.

FIG. 9 illustrates a basic block diagram for implementing the real-timetime measurement method of the present invention.

FIG. 10 depicts an embodiment of a sampling system, according to someembodiments of the present invention.

FIG. 11 depicts an exemplary signal emanating from the summing node ofthe sampling system of FIG. 10.

FIG. 12 depicts a table that may be accessed to obtain a correctionfactor, according to some embodiments of the present invention.

FIG. 13 depicts a system for correction of distortion caused by thecombination of a probe, a front-end device, and circuitry within areal-time sampling device.

FIG. 14 depicts a step signal that is generated and measured by thesystem of FIG. 13.

DETAILED DESCRIPTION

Real Time Measurement System

Turning first to FIG. 9, an illustrative block diagram of a system thatcan be used to determine the time at which the input waveform 10 crossesthe amplitude A_(Threshold) is shown. The input waveform 10 and thethreshold amplitude A_(Threshold) 30 are applied to the time eventconversion block 20. The time event conversion block 20 generates anoutput event, such as a voltage transition, when the input amplitude 10crosses the amplitude threshold 30. The output of the time eventconversion block 20 is applied to the input to the limited slew eventcreation block 40. Upon receipt of the time event, the limited slewevent creation block 40 initiates an output event that includes anamplitude variation that is of fixed and repeatable amplitude andduration. The event's change in amplitude with respect to time, or slewrate, is limited so that the analog to digital (ADC) and dataacquisition block 50 can sample the event to determine a time thatcorresponds to the time that the event was initiated. The slew ratelimiting is effectively equivalent to bandwidth limiting for thepurposes of avoiding aliasing in the analog to digital conversionprocess. The data manipulation block 60 is used for collecting andmanipulating data.

It is important to note that the amplitude to time event conversionblock could be substituted with a block that converted some otherparameter to a time event. For example, optical polarity or some othertime varying parameter may be measured, thereby allowing this system tobe utilized in connection with measuring any number of time varyingparameters. More specifically, the time varying parameter could be amagnetic field. In this case, the magnetic field could be picked up witha magnetic loop antenna, amplified and applied to a limiting amplifier.The output of the amplifier would create the time event that is appliedto the limited slew event creation block 40. The resulting system wouldmeasure the time when the magnetic field strength exceeded a certainlevel.

Turning next to FIG. 1, a waveform to be measured 101 is presented tothe positive input of a comparator 102 at 100 while the negative inputto the comparator 101 is set to a voltage V_(Threshold) 103. The timingof the transition at the output of the comparator 101 is representativeof when the input waveform crossed the threshold amplitude. Given thatthe gain of the comparator function is high enough, and the comparatorcircuitry is chosen to have sufficient bandwidth, the step at the outputof the comparator 101 will have a very fast transition and settle timerelative to the step response of the succeeding filter 104. Thecomparator output must settle quickly if the output of the filter isexpected to settle quickly. Quick settle times ensure that energy from aprevious transition does not affect the current transition resulting inmeasurement error.

The output of the comparator 102 is applied to the input of a filter104. The characteristics of the filter 104 are chosen so that theresulting step response of the system has a very short settling time.The filter 104 also has the characteristics of limiting the bandwidth ofthe signal that exits the filter and is applied to the input of theanalog to digital conversion (ADC) and data collection block 105. Theaforementioned filter 104 may be embodied as a bandwidth-limitedoscilloscope. In other words, the bandwidth limitations of the circuitryof an oscilloscope may serve the function of the filter 104. This datacollection block can take the form of an assembly comprised of standardcircuits that are available on the market or it can be in the form of areal time sampling oscilloscope such as an Agilent Infiniium 80000series oscilloscope or a Tektronix TDS6000 series oscilloscope, as wellor any other suitable conversion technology. The data in the ADC anddata collection block 105 are accessed by the data manipulation block106 which operates on the data to extract the timing information. It ispossible to effectively double the sampling rate of the ADC and datacollection system by adding an additional ADC and data collection block108 in conjunction with the delay element 107. The delay is one half ofthe sampling period effectively interleaving sampling points. Theexample of doubling the sampling rate is given although any arbitrarynumber of additional channels can be added in order to increase theeffective sampling rate.

FIG. 2 at 200 illustrates an example input waveform 201 which is appliedto the comparator 102 of FIG. 1. The input waveform 201 crosses thestatic threshold voltage 202 at time T₀. For normal operatingtemperatures, the high and low amplitude values at the output of thecomparator 102 are stable and unchanging as are the propagation delaysof the comparator 102 and the filter 104 of FIG. 1. Still referring toFIG. 2, a resulting waveform transition 203 at the comparator output isinitiated a fixed time T₁ after the crossing of the input threshold 202.The amplitude of the comparator output transition is fixed at anamplitude “A”. At a fixed time T₂ later, the spread or stretched step204 is initiated at the output of the filter 104 in FIG. 1. Thiswaveform also has a fixed amplitude “A”, with the shape over time beingsimply the step response of the filter. A series of amplitudemeasurement samples are taken with the ADC and data acquisition block105 of FIG. 1. One such point is the point designated at 205 along thestretched step 204. This point has an amplitude and time associated withit (t₁,a₁) which is the result of the ADC amplitude measurement process.

It will be appreciated from FIG. 2 that to find the time whenV_(threshold) is crossed, one can work back from the point t₁,a₁ bysubtracting T_(lookup), T₁, and T₂ from the value of t₁ to get the valueT₀. In the case of jitter, the only values of interest are thedifferences between transition locations with the result being thatthere is no real sense of absolute time. Consequently, if a constanttime value were subtracted from all edge locations, such as T₁+T₂, thenthe time difference between edges would still remain the same. Theimplication of this in the context of locating T₀ is that all that needsto be subtracted from t₁ is T_(lookup). The value of T_(lookup) can befound if the characteristic shape of the expanded step 204 or referencewaveform is known. Assuming that the reference waveform 204 is known orcan be found, the lookup table 207 can be used to find the value ofT_(lookup) by identifying the ordinate value a₁ and locating thecorresponding or abscissa value T_(lookup). Subtracting T_(lookup) fromt₁ then produces a value representing the threshold crossing time.

One method for finding the reference waveform is to apply a clock likecalibration signal of known frequency to the input of the comparator102. The frequency of this calibration signal can be chosen to cause thesampling point 205 to appear to move slightly relative to the referencewaveform with each successive calibration signal edge. Given that thecalibration frequency is known, then a mathematical process ofoverlaying successive sampling points by subtracting the period of thecalibration signal from the appropriate points yields a very highresolution record of a single reference waveform. Averaging, smoothingand curve fitting are candidates for creating a model of the referencewaveform.

It will be appreciated in view of the disclosure, that even though thediscussion has described events associated with rising steps andtransitions, the described invention and embodiments also apply toevents of the opposite polarity. Further, there also exist otherreference waveforms that can be used to determine the times of thethreshold crossings. An alternative to the application of a step to thefilter is that of an impulse or a very short pulse. In this case theexpanded waveform would appear as a spread out impulse in contrast tothe step described above. Similar methods could be used for producingthe threshold crossing times.

Another method for measuring the transition times is described inconnection with FIG. 3. In this embodiment, a waveform is presented tothe positive input of a comparator 301 while the negative input to thecomparator is set to a voltage V_(Threshold) The output of thecomparator 301 is applied to the input of a one shot device 302 thatcreates a pulse of fixed width. The output of the one shot device 302 isapplied to the hold input of a hold function device 303. A referencewaveform such as a sinusoid 304 is applied to the input of the holdfunction device 303. The result at the output of the hold functiondevice 303 is illustrated at 305, namely a sinusoid output that is heldat a voltage V_(hold) corresponding to the point in time T_(hold) thatthe threshold transition occurred. This voltage is held for a period oftime corresponding to the pulse width of the pulse generated by the oneshot device 302. This output is applied to an analog to digitalconversion (ADC) and data collection block 306 similar to the onedescribed above.

Given the appropriate choice of reference frequency and sampling rate, aprescribed number of sample points will exist on the “held” portion ofthe output of the hold function device 303 (as shown at 305). If theshape of the reference waveform is well known, as in the case of a wellfiltered sinusoid, the location of the threshold crossing can beinferred by the held voltage. The phase of the sinusoid can beestablished by applying the sinusoid to another ADC and data acquisitionelement 309. It may be beneficial to have the output of the one shotdrive two hold function devices 303 and 307. In this case the input tothe hold functions would be reference sinusoidal waveforms that are outof phase by 90 degrees which is caused by the phase shift element 310.This would insure that one of the hold functions would be holding avoltage of sufficient slew rate as to accurately imply the thresholdcrossing. This implementation has the advantage of not relying on theaccuracy of the sampling time base.

The output from the second hold function device 301 is provided to ADCand Data Acquisition block 309, while a data manipulation block 311accepts the data from ADC and data acquisition blocks 306, 308 and 309.

FIG. 4 illustrates an embodiment with two divider blocks 402 and 404inserted into the measurement system. The non-inverting output of thecomparator 401 is applied to the input of a first divider block 402.Assuming the dividers are positive edge triggered, then all positivetransitions applied to the comparator 401 will cause a positive edge tobe applied to the first divider 402. Since the divider block 402 ispositive edge triggered, the output will only transition in response toall positive edges. The successive measurement circuit 403 willtherefore only be able to respond to events with positive transitions.Conversely, the inverting output of the comparator 401 only transitionshigh in response to negative going threshold crossings at its input. Thedivider block 404 and the subsequent measurement channel thus onlyrespond to negative going threshold crossings. The divider blocks 402and 404 perform the function of cutting the frequency of edges enteringthe measurement channels 403 and 405 in half. This presents two options.First, if the hardware that would have been used for a singlemeasurement channel is duplicated, then the system can be made to run attwice the data rate of a single channel system. The second option is,for a given data rate, that the number of measurement channels can bedoubled, and the filter frequency, and the ADC and Data Acquisitionrate, can be reduced by half. This can result in reduced cost since thecost of the ADC and Data Acquisition block may drop non-linearly. It isimportant to note that more than two dividers can be employed, therebyallowing a wide variety of configurations with regard to number ofmeasurement channels. This permits the architecture to be very scalable.

An Oscilloscope Architecture Based on the Real Time MeasurementCapability

Conventional equivalent time oscilloscopes depend on precision delaycircuitry to create precise trigger locations for sampling waveforms.Precision for longer delay values is very difficult to achieve and therate at which the trigger delay can be recycled can also be ameasurement rate limitation. With the new time measurement capabilitiesof the present invention, the architecture shown in FIG. 5 can beimplemented. Such architecture not only measures the sampled voltage,but also measures the time at which the sampling trigger occurred.

In FIG. 5, a waveform 501 to be measured is applied to the input of ahold function block 502. The hold function block 502 holds the incomingwaveform 501 for a prescribed amount of time. The ADC and dataacquisition block 504 samples the held voltage yielding the amplitude ofthe input to the hold function at the time of the trigger event. Theoutput of the trigger generation circuit 503 is also applied to theinput of the time measurement circuitry block 505 which measures theexact time of the trigger event. The long range accuracy of the timemeasurement method is far superior to the delay generation capability ofcurrent equivalent time oscilloscopes, thus providing greatly improvedtiming accuracy. The measurement rate afforded by the new timemeasurement block 505 removes the limitation of the delay recycle timethat is imposed by current equivalent time oscilloscopes allowingmeasurement rates to be increased dramatically. The output is providedto data manipulation block 506.

The embodiment illustrated in FIG. 6A utilizes this approach and isexemplary for measuring eye diagrams at a rate that is much higher thanpreviously possible and without the need for a trigger signal. It alsoprovides performance levels equal to or better than those of theequivalent time scope such as bandwidth, amplitude noise floor, jitternoise floor etc.

The configuration of this embodiment is similar to that shown in FIG. 5with the block 606 comprising the various blocks illustrated in FIG. 5,with the exception of the trigger generation block 503. Instead, arandomized trigger source is illustrated in FIG. 6 generally at 601. Thestart up circuitry 604 initiates an edge on the one shot device 602which is applied to the delay element 603. The propagation delay of thedelay element 603 is a voltage controlled delay element. A random noisevoltage source 605 is applied to the delay element 603 causing the delayto vary randomly over time. The output of the delay is applied to theamplitude and time acquisition system 606, and the one shot 602,retriggering the loop. The variation the delay element 603 is set so asto keep successive triggers from being time correlated. This causes thesamples that are taken to be randomly spaced in time ensuring that agood distribution of times are sampled relative to the data rate of thesignal. To that end the variation in delay should be a substantialportion of a bit period RMS.

Performance enhancement can be further achieved in several ways. Forexample, referring to FIG. 5, if the hold time of the track and holdblock 502 is extended, then multiple samples of the voltage that isbeing held can be taken by the ADC and data acquisition block 504. Thesamples can then be averaged by the data manipulation block 506, whichreduces the noise floor of the resulting amplitude measurement.

An alternative embodiment of a randomized trigger source is depicted inFIG. 6B.

A circuit for reducing the time measurement error is shown in FIG. 7.Here, a single event represented by a transition 701 is applied to theinput of the logical “or” block 702. Assuming that the other input isnot asserted, then the output of the “or” block 702 transitions. Thistransition is applied to the input of the one shot device 703. The oneshot device 703 creates a pulse of fixed width that is applied to theinput of the delay element 704. The output of the delay element 704 isapplied to three other elements. First it is applied to the timemeasurement circuitry 705 which measures the time of this transitionutilizing the methods previously described. The output of the delayelement 704 is also applied to the logical “or” block 702 causing theone shot to once again trigger. The output of the delay element 704 isalso applied to the loop counter element 706 which counts the number ofpasses the loop has propagated. When the count reaches a terminal countthe terminal count signal is asserted. The terminal count signal is thenapplied to the reset logic 707. The reset logic then asserts the resetinputs on the loop counter 706 and the one shot device 703, terminatingthe loop and resetting the counter 706.

The count value of the loop counter 706 can be programmable, and defineshow many cycles an event is allowed to circulate before the loop isterminated. If the number of cycles is N and the jitter of the loop issmall relative to the jitter of the time measurement circuitry 705, thenthe noise floor of the multiple pass measurement will drop the squareroot of N as compared to the single pass measurement.

Although the systems referred previously can be constructed out of anycomponents suitable to provide the described functionality, onepreferred configuration is described in FIG. 8. The test set 804contains circuitry that conditions and converts the input waveformssignals so that signals applied to a real time sampling oscilloscope801. The real time sampling oscilloscope incorporates the ADC and datafunctionality 805, 806, 807 functions as well as the data manipulationfunctionality 808. The real time oscilloscope processor and operatingsystem 809 runs a commonly available operating system (such the Windowsoperating system) and is capable of supporting other applications aswell. Software for the system control and applications can reside on thereal time oscilloscope processor system 809. The real time oscilloscopeprocessor 809 can also perform the task of communicating with the testset control logic and processor 810. This communication will occur oversuitable interconnect media 811 (e.g., an Ethernet connection).

FIG. 8 illustrates the test set 804 with differential signals 802, 803applied, but a single ended signal could be applied as well. The signals802, 803 are applied to splitting circuitry 812, 813 which split theinput signals and apply them to the inputs of time measurementconditioning circuitry 814 and the differential track and hold circuitry815 which is used for amplitude measurement. The output of the timemeasurement conditioning circuitry 814 exits the test set 804 and isapplied to the ADC and data acquisition circuitry 805 in the real timeoscilloscope 801 in order to perform the time measurement functiondescribed previously. The output of the time track and hold circuitry815 exits the test set 804 and is applied to the ADC and dataacquisition circuitry 806 in the real time oscilloscope 801 in order toperform the amplitude measurement function described previously. Thetrigger generation circuitry 816 creates the trigger events for thetrack and hold circuitry 815 which are also applied to the input of timemeasurement conditioning circuitry 817. The output of the timemeasurement conditioning circuitry 817 exits the test set 804 and isapplied to the ADC and data acquisition circuitry 807 in the real timeoscilloscope 801 in order to perform the time measurement functiondescribed previously. This configuration is capable of measuring theamplitude of the incoming differential waveforms and measuring the timeof transitions on the waveforms simultaneously.

As will be appreciated by those of skill in the art upon a review of thepresent disclosure, the embodiments described herein are one of manythat can be conceived to measure the timing and amplitude of waveformswith the concepts previously described for amplitude and timemeasurement.

The signal split 812 sends a signal out of the test set 804 which isapplied to the ADC and data acquisition circuitry 818 in the real timeoscilloscope 801. This signal can be used in conjunction with the realtime oscilloscopes typically elaborate trigger function 819 to initiatea measurement.

FIG. 10 depicts an alternative embodiment of the sampling method andsystem described herein. Briefly, the embodiment of FIG. 10 functions,so as to combine a timing signal with a signal to be sampled by areal-time sampling device, such as an accompanying real-timeoscilloscope. For the sake of discussion, the real-time sampling deviceis referred to herein as a real-time oscilloscope, although this neednot be the case, and the methods, systems, schemes, and techniquesdiscussed herein may work in conjunction with an real-time samplingdevice. The effect of this embodiment is to reduce the number of inputports occupied on the aforementioned oscilloscope. As can be seen fromFIG. 10, the relevant portion of the front-end device includes a pair oftrack and hold elements 1000 and 1002, respectively. The track and holdelements 1000 and 1002 are coupled to a differential input port of thefront-end device. The first track and hold element 1000 receives thepositive portion of the differential signal, and the second track andhold element 1002 receives the negative portion of the differentialsignal.

Each track and hold element 1000 and 1002 operates such that, upon thevoltage at its track input exceeding a threshold, the voltage exhibitedat its output tracks its input. The track input of each track and holdelement 1000 and 1002 is provided with a fixed pulse width signalyielded from a strobe generator 1004. The signal yielded therefrompasses through a variable delay element 1006 or 1008 as it propagates tothe track input of each track and hold element 1000 and 1002. The strobegenerator 1004 is configured to generate a clock signal that exhibitstrack pulses at random points in time (in this case, a “track pulse”refers to an occurrence in which the voltage of the signal emanatingfrom the strobe generator 1004 crosses the aforementioned thresholdimposed by the track and hold elements 1000 and 1002 for a fixed periodof time). Stated another way, the strobe generator 1004 generates trackpulse signals exhibiting fixed pulse width that occur at random pointsin time, relative to the unit interval of the differential signalapplied to the track and hold elements 1000 and 1002. The effect of suchrandomization is to ensure that, over the course of a sufficientlynumerous set of measurements, voltage samples are taken at substantiallyall points in time of the aforementioned unit interval. The strobegenerator 1004 is further configured to generate track pulses that aresubstantially identical in shape from track pulse to track pulses. Suchsubstantial repeatability of shape may be obtained, for example, by useof a high-speed logic configured to perform a one-shot type offunctionality.

As mentioned previously, variable delay elements 1006 and 1008 areinterposed between the strobe generator 1004 and the track inputs of thetrack and hold elements 1000 and 1002. The delay imposed by eachvariable delay element 1006 and 1008 may be adjusted to compensate forthe varying delays exhibited by the two track and hold elements 1000 and1002, i.e., the delay exhibited, for each track and hold element,between the point in time at which a logic level transition is exhibitedat its track input and the point in time at which the “holding” actionof each element 1000 and 1002 commences. Thus, if the first track andhold element 1000 imposes a delay, d₁, and the second element imposes adelay, d₂, the first variable delay element 1006 may be adjusted toimpose a delay of

d₂-d₁, assuming that d₂>d₁.

The outputs of the track and hold elements 1000 and 1002, and the clocksignal emitted by the strobe generator 1004 are combined by a summingnode 1010 (the input receiving the output of the second track and holdelement 1002 is negated). Accordingly, the output of the summing node1010 yields a signal having the clock signal emitted by the strobegenerator 1004 superimposed over the “held” signal to be measured by thereal-time oscilloscope. An example of the signal yielded by the summingnode 1010 is depicted in FIG. 11.

FIG. 11 depicts the output of the summing node 1010, immediatelypreceding and following a point in time when a clock signal emitted bythe strobe generator 1004 causes the track and hold elements 1000 and1002 to sample an incoming differential signal having a voltage of v₁.As described previously, the output of the summing node 1010 is coupledto an input port of a real-time oscilloscope. Consequently, the signalemitted therefrom is sampled by the real-time oscilloscope at points intime determined by a clock internal to the oscilloscope. The varioussamples taken by the real-time measurement device, such as anoscilloscope are represented as darkened dots in FIG. 11.

The signal depicted in FIG. 11 exhibits a rising edge 1102. Followingthe rising edge 1102, the signal plateaus, after which, it exhibits asubstantially constant voltage 1100 equal to the difference between thevoltages held by the track and hold devices 1000 and 1002. Thus, byobserving the voltage samples taken by the oscilloscope during theperiod of substantially constant amplitude 1100, the voltage level ofthe differential signal at a given sample time may be determined (e.g.,the voltage levels may be averaged to reduce the effects of noise).Providing an adjustable hold time allows the measurement rate to betraded off for increased resolution and reduced noise floor.

It should be noted that many oscilloscopes exhibit a fixed signal tonoise ratio. Such a fixed ratio may prove problematic when thedifference between the amplitudes of successive held signals isrelatively small, and the amplitude of the track pulse isdisproportionately large, because the signal variations in the heldsignals will be corrupted by noise. Thus, according to some embodiments,the amplitude of the track pulse is a function of the difference of theamplitude of successive held signals.

The track pulse is of a fixed duration, and the rising edge 1102 may beused to mate the aforementioned voltage level 1100 exhibited by thedifferential signal with a sample time. It is known that the hold timeoccurs at some point in time on the falling edge of the track pulse(this is because the sample and hold elements 1000 and 1002 operate, soas to perform their respective “hold” operations when the voltage attheir respective track inputs falls beneath some threshold). Because theshape of the track pulse emitted from the strobe generator issubstantially identical with each pulse, the hold time may be related toa sample obtained by the real-time oscilloscope, occurring on the risingedge 1102 of the track pulse. Thus, for example, the sample identifiedby reference numeral 1104 may be identified as occurring on the risingedge 1102 of the track pulse. After identifying this sample 1104 asoccurring on the rising edge 1102 of the track pulse, its amplituderelative to the baseline voltage preceding the rising edge 1102 isobtained. This amplitude is termed a_(baseline). Then, because the shapeof the track pulse is known, the amplitude of any point on the risingedge 1102 may be related to a correction factor that indicates theamount of time by which that point (e.g., point 1104) either precedes orsucceeds the sampling time. Accordingly, as shown in FIG. 12, a tablerelating the aforementioned value, a_(baseline), to a correction factormay be accessed, and a particular correction factor associated with thevalue of a_(baseline) may be retrieved (interpolation may be used, ifneeded). Then, the sample time may be arrived at by the followingequation:t _(sample) =n*rate_(sampling)+fact_(correction),where n represents the sample number of the selected point on the risingedge of clock pulse (e.g., the selected point on the rising edge was then=1067th sample in the sequence), rate_(sampling) represents thesampling rate, and factor_(correction) represents the aforementionedcorrection factor obtained from a table, such as the one described withreference to FIG. 12. (Of course t_(sample), may be arrived at by use ofa function or curve-fit procedure that generates values similar to thosepopulating the table of FIG. 12.)

FIG. 13 depicts a system for obtaining a transfer function imposed bythe combination of the particular probe used to interface the front-enddevice with a device under test, the front-end device, and the circuitryof the oscilloscope. It should be noted that many probes are configuredto provide configuration data to the oscilloscope with which they arecoupled. The configuration data informs the oscilloscope of the transferfunction imposed by the probe, so that the oscilloscope cansubstantially correct any distortion created by the probe. The followingscheme permits the front-end device disclosed herein to function with aprobe having an unknown transfer function, and does not require thefront-end device to read any configuration data stream provided by theprobe, in order to determine the transfer function imposed thereby.

The system includes a probe 1300 for interfacing with the device undertest, and for obtaining the signal to be measured. The probe 1300interfaces with an input port 1302 provided by the front-end devicedescribed herein. The front-end device is provided with a conductive pad1304 located on a surface accessible by a user of the device. Theconductive pad 1304 is coupled to a step function generator 1306 (inprinciple, the step function generator 1306 can generate any knownfunction having wide band frequency content, but for the sake ofexample, the generator 1306 is described herein as a step functiongenerator). The step function generator 1306 may actually generate aslow square wave, which in the context of the data rates to be measuredby the front-end device and oscilloscope fulfills the role of a stepfunction. The generator 1306 may be composed of, for example, aprecision oscillator coupled to a high-performance comparator, so thatthe square wave/step function yielded thereby is of substantially thesame shape with each cycle.

At the time of manufacture, a time-sequenced set of voltages describingone cycle of the signal emitted by the step function generator 1306 maybe stored in a memory device within the front-end device. Thistime-sequenced set of voltages may be termed u(t). Alternatively oradditionally, a Fourier transform of u(t) may be stored in theaforementioned memory device. The transform of u(t) may be termed u(ω).

To obtain the transfer function imposed by the combination of the probe1300, the front-end device, and the circuitry of the oscilloscope, theprobe 1300 may be put into electrical contact with the conductive pad1304. Consequently, the combination formed by the front-end device andthe oscilloscope cooperate to measure the step function, u(t), using theaforementioned sampling technique discussed herein (in other words, themeasurement is built up, sample-by-sample, over the course of manysuccessive measurements, as is known in the art). Therefore, theoscilloscope obtains a time-sequenced set of voltages describing thestep function, as observed by the oscilloscope's measurement circuitry.This time-sequenced set of voltages is distorted by the transferfunction of the probe, front-end device, and measurement circuitry ofthe oscilloscope, meaning it is not equal to the time-sequenced set ofvoltages stored in the front-end device. (In other words u(t) andu_(m)(t) may differ, as shown in FIG. 14). The time-sequenced set ofvoltages describing the step function as measured by the oscilloscopemay be termed u_(m)(t). Again, the Fourier transform of u_(m)(t) may befound, and stored in the memory of the oscilloscope (or stored in anyother computing system in communication therewith, including thefront-end device). The transform of u_(m)(t) may be termed u_(m)(ω).

It is known that u(ω)·P(ω)·E(ω)=u_(m)(ω), where P(ω) represents, infrequency space, the transfer function imposed by the probe, and E(ω)represents, in frequency space, the transfer function imposed by thecombination of the front-end device and the circuitry of theoscilloscope. Therefore, P(ω)·E(ω)=u_(m)(ω)/u(ω)). Thus, the quantityarrived at by dividing u_(m)(ω) and u(ω) may be stored in memory of theoscilloscope or any other device in communication therewith, includingthe front-end device or another computer, and used to correctmeasurements. The correction herein may be performed using eitherspectral or time domain techniques.

For example, in the case of a signal X(t) having been measured using thesampling technique discussed herein (again, by “measured” it is meantthat the signal is built up, on a sample-by-sample basis over the courseof many iterations of the repeating pattern), the signal may becorrected by the aforementioned quantity, according to any of a numberof known techniques. X(t) may be transformed into frequency space,becoming X(ω). Then, X(ω) may be corrected by multiplication byu(ω)/u_(m)(ω). Thereafter, the resulting product may beinverse-transformed back into time space, and the results may bedisplayed by the oscilloscope, or otherwise used in the normal mannersthat oscilloscopes use sample data.

It is to be noted that any of the foregoing data manipulation schemesdescribed with reference to any of the foregoing Figures, includingthose described with reference to Figures with reference to FIGS. 10-14may be stored as a set of instructions in a memory device or any form ofcomputer-readable medium. The instructions may be executed by anyprocessor having access to the data to be manipulated, including,without limitation, a processor embedded in an oscilloscope mated to thefront-end device described herein, a processor within the front-enddevice, or a computer in communication with either the oscilloscope orthe front-end device.

It will be appreciated that the principles of this invention apply notonly to devices for measuring time voltage threshold events, but also toother environments in which the measurement of time varying parametersand functions is desirable. While particular embodiments of theinvention have been described with respect to its application, it willbe understood by those skilled in the art that the invention is notlimited by such application or embodiment or the particular componentsdisclosed and described herein. It will be appreciated by those skilledin the art that other components that embody the principles of thisinvention and other applications therefor other than as described hereincan be configured within the spirit and intent of this invention. Thearrangements described herein are provided as examples of embodimentsthat incorporate and practice the principles of this invention. Othermodifications and alterations are well within the knowledge of thoseskilled in the art.

1. A method of sampling a signal, the method comprising: providing thesignal to a track and hold element; causing the track and hold elementto perform a hold operation, thereby yielding a held signal exhibiting asubstantially constant voltage level, wherein the act of causing thehold operation includes providing a track signal having a substantiallypredetermined shape to the track and hold element; providing the heldsignal to a signal measurement device; providing the track signal to thesignal measurement device, so that the signal measurement device samplesthe track signal, and obtains at least one sample on an edge of thetrack signal; and using the at least one sample and the predeterminedshape of the track signal to determine a time value associated with theheld signal.
 2. The method of claim 1, wherein the signal measurementdevice associates the at least one sample of the edge of the tracksignal with a point in time, relative to a start of a data record, atwhich the sample was obtained, and wherein the act of using the at leastone sample and the predetermined shape of the track signal to determinea time value associated with the held signal comprises: accessing atable to obtain an offset value, based upon the at least one sample ofthe edge of the track signal; and adding the offset value to the pointin time associated with the sample of the edge of the track signal. 3.The method of claim 2, wherein the table is populated with offset valuesdetermined based upon the predetermined shape of the track signal. 4.The method of claim 3, wherein the signal measurement device comprisesan oscilloscope.
 5. The method of claim 1, wherein the signalmeasurement device comprises an oscilloscope.
 6. The method of claim 1,wherein the act of causing the track and hold element to perform a holdoperation is performed a plurality of times, so as to obtain a pluralityof samples of the signal, and wherein the plural acts of causing thetrack and hold element to perform a hold operation occur at randompoints in time.
 7. The method of claim 6, wherein a measure of centraltendency of times between hold operations is selectable.
 8. The methodof claim 2, wherein the track signal and the held signal are addedtogether, yielding a summed signal.
 9. The method of claim 8, whereinthe acts of providing the held signal and track signal to the signalmeasurement device comprises providing the summed signal to the signalmeasurement device.
 10. The method of claim 1, wherein the signal is adifferential signal, and wherein a single analog input is provided tothe signal measurement device, the single analog output including bothpolarities of the differential signal and the track signal.
 11. Themethod of claim 1, wherein an amplitude of the track signal is afunction of a difference between amplitudes of successive held signals.